Plate 2 Should Be Re Done
Errata Sheet for Volume 2 of Practical RF Circuit Design for Modern Wireless Systems
Page 23: Figure 1.13 The open-loop gain should be:
(s)
G
(s) instead of G
OL
OL
Page 67: Figure 1.54 The various base current values are missing in the transfer characteristics plot.
They are 25µA/step.
Page 185 Labels hard to decipher – see correct figure:
Page 240: delete 2 in equation i.e. V
= V
PEAK
DD
Page 245 Figure 5.18
Figure is from ‘Solid State Radio Engineering’, Figure 12-4,
Krauss, Bostian, and Raab, J. Wiley and Sons, New York, 1980.
Page 251 Figure 5.22
Figure is from ‘Solid State Radio Engineering’, Figure 12-6,
Krauss, Bostian, and Raab, J. Wiley and Sons, New York, 1980.
Page 260 Figure 5.28
Figure is from ‘Solid State Radio Engineering’, Figure 14-9,
Krauss, Bostian, and Raab, J. Wiley and Sons, New York, 1980.
Figure is from ‘Solid State Radio Engineering’, Krauss, Bostian, and Raab, J. Wiley and Sons,
New York, 1980.
Page 262
Figure 5.29
Figure is from ‘Solid State Radio Engineering’, Figure 14-10,
Krauss, Bostian, and Raab, J. Wiley and Sons, New York, 1980.
Page 389-391: Replace text with the following (emitter resistor R1 was disconnected in original
simulation):
A linear analysis of the impedance seen looking into the ‘device’ is shown in Figure 6.43. The
Smith chart confirms that its impedance lies outside the unit circle with a reflection coefficient
greater than one (i.e. the input impedance contains a negative real part). The real and imaginary
values of this impedance are plotted in Figure 6.44, together with the total magnitude of the
impedance.
INDQ
ID=L1
RES
L=1e5 nH
Q=0
PORT
ID=R4
FQ=0 GHz
P=1
R=100 Ohm
ALPH=1
Z=50 Ohm
S11
RES
CAP
CAP
Swp Ma x
ID=R3
ID=C7
ID=C6
2GHz
R=5000 Ohm
C=4.7e6 pF
C=10 pF
S(1,1)
2 C
800 MHz VCO circuit
5
5
SUBCKT
1
0.
1.0
0.
ID=S1
NET="BFQ67"
B
2 .
3 E
0
2.0
RES
RES
CAP
CAP
CAP
CAP
ID=R2
ID=R1
ID=C5
ID=C2
ID=C3
ID=C4
R=1500 Ohm
R=470 Ohm
C=1e6 pF
C=10 pF
C=1 pF
C=1.8 pF
5
0
0
.
0
.
.
-0
0
1.
-3
-2
-2.0
- 0. 2
-1.0
5
.
.5
-0
-0
S wp Min
0.1 GH z
Figure 6.43. (a) The 'device' model
for the oscillator of Figure 6.42, and
(b) the impedance looking into the
'device' output
Device impedance
0
3000
0.8 GHz
-76.1
-1000
2000
0.8 GHz
-313.5
Magnitude of impedance (R)
800 MHz VCO circuit
-2000
Re(Z(1,1)) (L)
1000
800 MHz VCO circuit
Im(Z(1,1)) (L)
800 MHz VCO circuit
-3000
0
0.1
0.3
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9 2
Frequency (GHz)
Figure 6.44. The impedance seen at the
output port of the 'device' in Figure 6.43,
showing the magnitude and its real and
imaginary parts.
As anticipated for the Colpitts configuration, a very broadband negative resistance is achieved
between at least 100MHz and 2GHz. This confirms the usefulness of this topology. At 800MHz,
we see that for steady-state oscillation the resonator loading the device will need to provide a
resistive load smaller than 76-ohms in series with an inductance of +j313 ohms.
Figure 6.45 shows the nature of the resonant ‘load’ that is connected to the collector of the Clapp
oscillator. The circuit topology looks suspiciously like a shunt R-L network, when in fact the
Colpitts topology requires a series R-L circuit as its load to achieve a 90-degree crossing angle
and to avoid multiple oscillatory modes. However, the impedance plot shows that across this
range of frequencies the impedance lies on a line of almost constant resistance, so that the
oscillator is indeed correctly terminated with a series circuit. Depending on the value of the
varactor capacitance used, the resonant circuit can be made inductive at frequencies above about
750MHz, in series with a resistance that is less than the required 76 ohms. The load does in fact
also pass through a shunt resonance around 836MHz, where the quarter-wave line appears as an
open circuit. However, because the circuit is used below this frequency in its inductive region, in
much the same way as a crystal is used at lower frequencies below its parallel resonance, there is
the possibility of only a single oscillation frequency. The load has an inductance of +j313 ohms
at just one frequency, depending on the value of the varactor tuning. Here, we have adjusted the
varactor capacitance so it occurs at 800MHz exactly.
Input reflection coefficient
0
8
Swp Max
.
1.
0
6
850MHz
. 0
2. 0
4 . 0
CAP
CAP
S(1,1)
PORT
3. 0
ID=Cx
ID=C2
P=1
C=0.5 pF
C=1 pF
4. 0
Z=50 Ohm
5. 0
2
. 0
800 MHz
r 7.86 Ohm
10. 0
TLSC
x 313 Ohm
CAPQ
0
2
4
6
8
0
0
0
0
0
ID=TL1
ID=Cvaractor
0
0.
0.
0.
0.
1.
2.
3.
4.
5.
10.
Z0=50 Ohm
C=2.672 pF
EL=90 Deg
Q=40
F0=950 MHz
0
. 0
1
FQ=500 MHz
-
ALPH=1
- 0. 2
0
. 5 -
0 . 4 -
0 . 3 -
- 0. 4
0 . 2
6
-
.0-
8.
0
Swp Min
0
.
-
-1
750MHz
Figure 6.45. (a) The resonant circuit of the Clapp oscillator of Figure 6.42, loading
the 'device' of Figure 6.43. (b) the input reflection coefficient of the resonator circuit
This is illustrated in more detail in Figure 6.46, which shows the impedance variation of the
resonator around 800MHz. As the varactor capacitor is tuned between 1pF and 3pF, the
frequency at which the inductance of the resonator cancels out the net capacitance of the Clapp
‘device’ (-j313 ohms) changes from 829 MHz to 797 MHz. With the varactor capacitance set at
2.672pF as shown, the oscillation frequency will be exactly 800MHz. We can calculate the
reactance slope with frequency using values 1MHz either side of this oscillation frequency,
where the reactance changes 41 ohms over a 2MHz frequency increment. We can then use the
equation (6.34) for Q to estimate
6
f
dX
800 *10
41
0
Q =
=
=
(6.44)
L
2 R
df
2
D
(76)
108
6
2 *10
f0
This expression uses the oscillator model of Figure 6.14. The reactance slope in the above
expression is set by the resonant circuit itself, whose X
∆ is dominant. The value of device
resistance to use is difficult to estimate, but we have taken the small-signal value from Figure
6.44 which is its largest possible magnitude. As the coupling - or step-up - capacitor Cx is
decreased, the reactance slope seen by the device at resonance increases, and the oscillator
loaded Q also increases. Varying Cx is a very effective way of changing the relationship between
the loaded Q and external Q, and thus trading off output power and phase noise.
Tuning impedance
30000
836 MHz
800 MHz
-0
20000
313
801 MHz
799 MHz
334
10000
293
0
-10000
Im(Z(1,1))
-20000
Oscillator load
-30000
750
800
850
Frequency (MHz)
Figure 6.46. The reactance of the resonant load of the circuit of Figure 6.42
around the oscillation frequency
Page 549: Under the heading, “Ideal lumped inductance and capacitance…” the correct capacitances
are:
159
159B
C
C
=
=
pF
fGHzXC
fGHz
Note: The quantities 0.159 and 159 need to be interchanged from their printed values.
Under the heading, “Ideal stub reactance, susceptance, and electrical length,” in the “Open stubs”
section, the signs of the two “j” operators should be interchanged:
In the first line, the [+j] should read [-j]
In the second line [-j] should read [+j]
Note: The signs under “Shorted stubs” are correct
